User-definable thermal drift voltage control oscillator

ABSTRACT

A voltage controlled oscillator that includes a slot-cut-microstrip-line coupled between a resonator, a tuning diode network and an active device and being operable to act as a common-coupling capacitor between the resonator, the tuning diode network and the active device.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of the filing date of U.S.Provisional Patent Application No. 60/527,957 filed Dec. 9, 2003;60/528,670 filed Dec. 11, 2003; and 60/563,481 filed Apr. 19, 2004, thedisclosures of which are hereby incorporated herein by reference.

BACKGROUND OF THE INVENTION

A voltage controlled oscillator (VCO) or oscillator is a component thatcan be used to translate DC voltage into a radio frequency (RF) voltageor signal. The magnitude of the output signal is dependent on the designof the VCO circuit and the frequency of operation is determined by aresonator that provides an input signal. Clock generation and clockrecovery circuits typically use VCOs within a phase locked loop (PLL) toeither generate a clock from an external reference or from an incomingdata stream. VCOs affect the performance of PLLs. In addition, PLLs aretypically considered essential components in communication networking asthe generated clock signal is typically used to either transmit orrecover the underlying service information so that the information canbe used for its intended purpose. PLLs are also important in wirelessnetworks as they enable the communications equipment to quickly lockonto the carrier frequency on which communications are transmitted.

The popularity of mobile telephones has renewed interest in andgenerated more attention in wireless architectures. This popularity hasfurther spawned renewed interest in the design of low noise widebandoscillators. The recent explosive growth in the new families of cellulartelephones and base stations using universal mobile telephone systems(UMTS) has stirred a need for developing an ultra-low noise oscillatorwith a fairly wide tuning range. The demands of wideband sources havegenerally increased telescopically because of the explosive growth ofwireless communications. In particular, modern communication systems aretypically multi-band and multi-mode, therefore requiring a wideband lownoise source that preferably allows simultaneous access to DCS 1800, PCS1900 and WCDMA (wideband code division multiple access) networks by asingle wideband VCO.

The dynamic operating range and noise performance of a VCO may limit oraffect the performance of the PLL itself, which in turn may affect theperformance of the device in which the PLL is employed, e.g., RFtransceivers, a cell phone, a modem card, etc. Broadband tunability ofVCOs represents one of the more fundamental tradeoffs in the design of aVCO, impacting both the technology and the topology used. The dynamictime average quality factor (i.e., Q-factor) of the resonator as well asthe tuning diode noise contribution affect the noise performance of aVCO. Furthermore, the dynamic loaded Q is, in general, inverselyproportional to the operating frequency range of the VCO.

Despite the continuous improvement in VCO technology, low phase noisetypically remains a bottleneck and poses a challenge to RF transceiver(transmitter-receivers) design. In addition, oscillator/VCO designtypically poses a challenge to the RF trans-receiver system. This istypically considered due to the more demanding parameters of the VCOdesign: low phase noise, low power consumption and wide frequency tuningrange.

Improvements in oscillator/VCO technology have continued with time,yielding ever-smaller sources with enhanced phase noise and tuninglinearity but the phenomena of the thermal drift over the temperaturerange (−40° C. to +85° C.) has not been properly addressed. The wideoperating temperature range of the oscillator/VCOs coupled with ageneral lack of information on the thermal drift-profile creates a needfor the development of a uniform and user-definable thermal driftprofile oscillator with a relatively low thermal drift over the wideoperating temperature range and operating frequency band.

Usually, high-stability oscillators are built with a quartz crystal upto frequencies of several hundred megahertz. However, in order toachieve better stability and lower costs, the SAW (surface acousticwave) resonator based oscillator is generally considered a better choicefor an ultra low phase noise low thermal drift oscillator. SAWresonators are typically used in oscillators as a two-port resonator andhave a relatively small pull-in range that usually does not support asufficient tuning range to compensate for tolerances due to the circuitcomponents and thermal drift over the operating temperature range (−40°C. to +85° C.). In addition, SAW devices are comparatively expensivecompared to CROs (ceramic resonator based oscillator) and theiravailability and performance are limited to a selected frequency andnarrow operating temperature range (−20° C. to +70° C.) making themunsuitable for operating in stringent temperature environments and/orlow cost applications.

In addition, the thermal drift of a ceramic resonator basedoscillator/VCOs is typically around 5-10 MHz over a temperature range of−40° C. to +85° C. The ceramic resonator based VCO is usually alsosusceptible to phase hits that may occur in a PLL.

Thus, there is a need for a user-definable thermal drift oscillatoroperable over a wide temperature range, which offers a cost-effectivesolution to the phase hit problem.

SUMMARY OF THE INVENTION

An aspect of the present invention is an oscillator. The oscillatorpreferably comprises an active device having first, second and thirdterminals and circuitry coupled between the first and second terminalsof the active device. The circuitry is preferably operative to provide abias voltage to the active device and feedback a select amount of phasenoise to the active device.

The oscillator further preferably comprises a tuning diode coupled tothe second terminal of the active device through aslot-cut-printed-board coupling network.

In accordance with this aspect of the present invention, theslot-cut-printed-board coupling network desirably acts as an evanescentmode buffer between a resonator coupled thereto and the active device.

Further in accordance with this aspect of the present invention, theslot-cut-printed-board coupling network operates to control a profile ofthe thermal drift of the active device or, in general, the oscillator.

Further still in accordance with this aspect of the present invention, afeedback capacitor is preferably coupled between the second and thirdterminals of the active device. In addition, the oscillator may furtherdesirably comprise a first filter and a second filter coupled to thethird terminal so as to provide two-stage regenerative filter.

Further in accordance with this aspect of the present invention, theactive device may comprise a bipolar transistor or a field effecttransistor and the first, second and third terminals respectivelycomprise the collector, base and emitter nodes of either of thetransistors.

Another aspect of the present invention is an oscillator that preferablycomprises an active device and circuitry coupled between a resonator, atuning diode network and the active device. The circuitry is preferablyoperable to act as a common-coupling capacitor between the resonator,the tuning diode network and the active device.

In accordance with this aspect of the present invention, the circuitrycontrols a thermal drift profile of the oscillator over an operatingtemperature range. In accordance with this aspect of the presentinvention, the circuitry desirably comprises a slot-cut-microstrip-line,whose dimensions are selectable to define a thermal profile of theoscillator.

Further in accordance with this aspect of the present invention, thecircuitry acts as an evanescent-mode-buffer between the resonator andthe active device. Further still, the tuning diode network iscapacitively coupled to the circuitry.

Further in accordance with this aspect of the present invention, theresonator preferably comprises a ceramic resonator. Further still, theactive device desirably comprises either a field effect transistor or abipolar transistor.

In another aspect, the present invention comprises an apparatuscomprising a phase lock loop for generating a clock signal used totransmit or recover information communicated from or to the apparatus.In addition, the phase lock loop preferably comprises avoltage-controlled oscillator for generating the clock signal. Mostpreferably, the voltage-controlled oscillator preferably comprises anactive device; and a slot-cut-microstrip-line coupled between aresonator, a tuning diode network and the active device that is operableto act as a common-coupling capacitor between the resonator, the tuningdiode network and said active device.

Preferably, the apparatus comprises a wireless device and mostpreferably comprises a cellular telephone. In addition, the apparatusmay also comprise a personal digital assistant.

In another aspect, the present invention comprises an apparatus thatcomprises a phase lock loop for generating a clock signal used totransmit or recover information communicated from or to the apparatus.The phase lock loop desirably includes a voltage-controlled oscillatorfor generating the clock signal. The voltage controlled oscillatorpreferably comprises an active device; and a slot-cut-microstrip-linecoupled between a resonator, a tuning diode network and the activedevice and operable to act as a common-coupling capacitor between theresonator, the tuning diode network and said active device. Theapparatus may desirably comprise a wireless device, and most desirablycomprises a cellular telephone. Further in accordance with this aspectof the present invention, the apparatus preferably comprises a personaldigital assistant.

In another aspect, the present invention comprises a telephone. Thetelephone preferably comprises a phase lock loop for generating a clocksignal used to transmit or recover information communicated from or tothe telephone. The phase lock loop preferably comprises avoltage-controlled oscillator for generating the clock signal, thevoltage controlled oscillator comprising, an active device; andcircuitry coupled between a resonator, a tuning diode network and theactive device and operable to act as a common-coupling capacitor betweenthe resonator, the tuning diode network and said active device. Inaccordance with this aspect of the present invention, the informationmay be communicated over a wireless or wired network.

In a method aspect, the present invention comprises coupling a capacitorbetween a resonator, a tuning diode network and an active device; andoperating the capacitor as an evanescent mode buffer between theresonator and the active device to compensate for drifts in an outputfrequency of the oscillator due to temperature changes.

The method may further desirably comprise biasing the active device at apredetermined voltage such that the capacitor maintains thepredetermined voltage level by compensating for drifts in an outputfrequency of the oscillator due to temperature changes. Further inaccordance with the method, the evanescent mode buffer compensates fordrifts in the output frequency of the oscillator due to temperaturechanges by storing additional energy generated by the oscillator due totemperature changes.

BRIEF DESCRIPTION OF THE DRAWINGS

FIGS. 1A and 1B depict circuits of a tuning diode in accordance with anaspect of the present invention.

FIG. 2 depicts a schematic of an oscillator in accordance with an aspectof the present invention.

FIG. 3 depicts a schematic of an oscillator in accordance with an aspectof the present invention.

FIG. 4 depicts a schematic of an oscillator in accordance with an aspectof the present invention.

FIG. 5 illustratively depicts a user-definable thermal profile plot overthe temperature range of −40° C. to +85° C. in accordance with an aspectof the present invention.

FIG. 6 illustratively depicts a user-definable thermal profile plot overthe temperature range of −40° C. to +85° C. in accordance with an aspectof the present invention.

FIG. 7 illustratively depicts a user-definable thermal profile plot overthe temperature range of −40° C. to +85° C. in accordance with an aspectof the present invention.

FIG. 8 illustratively depicts a user-definable thermal profile plot overthe temperature range of −40° C. to +85° C. in accordance with an aspectof the present invention.

FIG. 9 depicts a schematic of an oscillator in accordance with an aspectof the present invention.

FIG. 10 illustratively depicts a phase noise plot of an oscillatorimplemented in accordance with an aspect of the present invention.

DETAILED DESCRIPTION

FIGS. 1A and 1B depict circuits that illustrate a tuning diode usingresistors, capacitors and inductors. As shown in FIG. 1A, a tuning diodemay be depicted as a two-port device (as shown, ports 1 and 2) having aresistor R_(s) connected to port 1 and in series with an inductor L_(s).R_(s) and L_(s) are connected in series to resistor R_(p) and a variablecapacitor C_(j), which are in parallel with each other. C_(j) reflectsthe junction capacitance of the tuning diode and is variable in responseto temperature changes. The circuit further includes a capacitor C_(c)in parallel with R_(s), L_(s) and C_(j) between ports 1 and 2 and aninductor L_(s) between port 2, C_(j), C_(c) and R_(p), as shown.

FIG. 1B shows a simplified equivalent circuit of a tuning diode andincludes resistor R_(p) in parallel with capacitor C_(c). The capacitorC_(c) is also in series with resistor R_(s).

With reference to FIGS. 1A and 1B, the expression for the junctioncapacitance of the tuning diode under a reverse bias condition is givenby:${C_{j}( {ɛ_{r},d_{j},A,V} )} = {\frac{\mathbb{d}Q}{\mathbb{d}V} = {\frac{ɛ_{0}{ɛ_{r}(T)}A}{d_{j}} = \lbrack \frac{ɛ_{0}ɛ_{r}A}{\frac{\lbrack {2K_{s}{ɛ_{o}( {V_{bi} - V_{A}} )}} }{q}\frac{ ( {N_{A} + N_{D}} ) \rbrack^{1/2}}{N_{A}N_{D}}} \rbrack}}$

Under the abrupt junction assumption, the depletion region thickness,d_(j), is given by:$d_{j} = \lbrack {\frac{\lbrack {2K_{s}{ɛ_{o}( {V_{bi} - V_{A}} )}} }{q}\frac{ ( {N_{A} + N_{D}} ) \rbrack^{1/2}}{N_{A}N_{D}}} \rbrack$

Where, N_(D) and N_(A) are the donor and acceptor volume densities.V_(bi) is the built-in potential and is given by:$V_{bi} = {\lbrack \frac{kT}{q} \rbrack{\ln\lbrack \frac{N_{A}N_{D}}{n_{i}^{2}} \rbrack}}$

Under reverse bias conditions, the spacing d_(j) is a function of theapplied voltage V_(A)<0 and this effect is used to produce a variablecapacitor. The equivalent capacitance of a junction per unit area isgiven as:${c( {q,B,ɛ} )} = {\frac{C_{j}( {ɛ_{r},d_{j},A} )}{A} = {{ɛ_{0}{ɛ_{r}(T)}\frac{\mathbb{d}E}{\mathbb{d}V}} = \lbrack \frac{{qB}\quad ɛ^{({m + 1})}}{( {m + 2} )( {V + \phi} )} \rbrack^{1/{({m + 2})}}}}$${c( {q,B,ɛ} )} = \frac{ɛ_{o}ɛ_{r}}{\frac{\lbrack {2K_{s}{ɛ_{o}( {V_{bi} - V_{A}} )}} }{q}\frac{ ( {N_{A} + N_{D}} ) \rbrack^{1/2}}{N_{A}N_{D}}}$Q=ε ₀ε_(r)(T)E

Where,

-   -   Q=Charge per unit area    -   ε=ε₀ε_(r), ε_(r) 32 Dielectric constant    -   A=Device cross sectional area    -   d=Depletion layer width    -   c=Capacitance per unit area    -   m=Impurity exponent    -   q=Charge    -   B=Magnetic field    -   T=Temperature    -   V=Reverse voltage applied across the diode    -   E=Electric field

Combining all the constants terms together, including the area of thediode, into the constant, C_(d), the expression for capacitance is givenas:${C_{j}( {ɛ_{r},d_{j},A,V} )} = \frac{C_{d}}{( {V + \phi} )^{\gamma}}$ C _(d) =C ₀(φ)^(γ)C=C _(c) +C _(j)(ε_(r) ,d _(j) ,A,V)

Where,

-   -   γ=Capacitance exponent and depends on the doping geometry of the        diode. Its value varies from ⅓ to 2 for Si (silica) diode. The        value of γ for an abrupt junction diode is ½, but such diodes        have a limited tuning ratio. For wideband tunability, a hyper        abrupt junction diode is preferred, and value of γ is 1 or 2.    -   φ=The junction contact potential (0.7V for Si (silica))    -   C₀=Value of capacitance at zero voltage    -   C_(c)=Case capacitance    -   C_(j)=Junction capacitance

The tuning ratio (TR) is given by${TR} = {\frac{C_{j}( {V_{2} = V_{\min}} )}{C_{j}( {V_{1} = V_{\max}} )} = \lbrack \frac{( {V_{1} + \phi} )}{( {V_{2} + \phi} )} \rbrack^{\gamma}}$

The oscillator frequency varies proportionally to 1/{square root}C andfor the linear tuning range junction capacitance should vary as 1/V²(γ=2). The frequency ratio is given as the square root of the tuningratio TR

The Q of the tuning diode is a function of the reverse bias voltage,frequency and temperature. The expression for the Q of the tuning diodeis given by:$Q = {{2{\pi\lbrack \frac{{Stored} - {Energy}}{{Dissipated} - {energy}} \rbrack}} = \frac{\omega\quad{CR}_{p}^{2}}{R_{p} + R_{s} + {\omega^{2}C^{2}R_{s}R_{p}^{2}}}}$$C = {\lbrack {C_{c} + {C_{j}( {ɛ_{r},d_{j},A,V} )}} \rbrack = {C_{c} + \frac{C_{d}}{( {V + \phi} )^{\gamma}}}}$

The Q of the tuning diode falls off at high frequency due to the seriesbulk-resistance R_(s) and can be expressed as$\lbrack Q\rbrack_{{High} - {frequency}} =  {\lbrack \frac{\omega\quad{CR}_{p}^{2}}{R_{p} + R + {\omega^{2}C^{2}R_{s}R_{p}^{2}}} \rbrack_{\omega\operatorname{>>}}\quad \approx \lbrack \frac{\omega\quad{CR}_{p}^{2}}{\omega^{2}C^{2}R_{s}R_{p}} \rbrack}\Rightarrow{{\lbrack \frac{1}{\omega\quad{CR}_{s}} \rbrack\lbrack Q\rbrack}_{{High} - {frequency}} \propto \frac{1}{R_{s}}} $

The Q of the tuning diode falls off at low frequencies due to the backresistance of the reverse-biased diode R_(p) and can be expressed as:$\lbrack Q\rbrack_{{Low} - {frequency}} =  {\lbrack \frac{\omega\quad{CR}_{p}^{2}}{R_{p} + R_{s} + {\omega^{2}C^{2}{RR}_{p}^{2}}} \rbrack_{\omega\operatorname{<<}} \approx \lbrack \frac{\omega\quad{CR}_{p}^{2}}{R_{p} + R_{s}} \rbrack}\Rightarrow{\omega\quad{CR}_{p}} $ [Q]_(Low-frequency)∝R_(p)

Where

-   -   R_(p)=Parallel resistance or back resistance of the diode    -   R_(s)=Bulk resistance of the diode-device material    -   L_(s)=Internal lead inductance    -   L_(s′)=External lead inductance    -   C_(c)=Case capacitance

As the junction-temperature increases, the leakage current increases andit lowers the back resistance R_(p) of the diode. The increase in thejunction temperature causes a slight decrease in R_(s), but the effectsof the decreasing R_(p) are greater and this forces the effective Q todecrease.

The change in the value of the capacitance of the tuning diode withrespect to temperature causes frequency drifts of the oscillator/VCOscircuit. The change in the value of the capacitance with temperature canbe given by: C ∝ [T]^(T_(cc))${C(V)} = \frac{C(0)}{( {V + \phi} )^{\gamma}}$$\frac{\mathbb{d}{C(V)}}{\mathbb{d}T} = {\frac{\gamma\quad{C(0)}}{( {V + \phi} )^{({\gamma + 1})}}\frac{\mathbb{d}\phi}{\mathbb{d}T}}$$T_{CC} = {{\lbrack \frac{1}{C(V)} \rbrack\lbrack \frac{\mathbb{d}{C(V)}}{\mathbb{d}T} \rbrack} = {- {\lbrack \frac{\gamma}{( {V + \phi} )} \rbrack\lbrack \frac{\mathbb{d}\phi}{\mathbb{d}T} \rbrack}}}$${\frac{\mathbb{d}\phi}{\mathbb{d}T} \approx {{- 2.3}( {{mV}/{\,^{0}C}} )}},{{for}\quad{{Si}({silica})}}$Where T_(cc) is a temperature coefficient.

From above, the temperature coefficient T_(cc) is inversely proportionalto the applied voltage and directly proportional to the diode slope γ.In addition, tuning diode capacitance increases with an increase intemperature, whereas capacitance drift decreases with an increase inreverse bias voltage, i.e., at a higher reverse voltage drift is at aminimum as compared to at a low reverse voltage. The capacitanceconstant C_(d) is a function of the geometric dimension and varies withthe dielectric constant, which is also a function of temperature.

The net thermal drift of an oscillator/VCO is generally due to thetuning diode, active device, resonator and passive components in theoscillator circuitry. The approach of adding a negative temperaturecoefficient compensating capacitor typically does not compensate for thetuning diode temperature coefficient T_(cc) because the change in thecapacitance is not constant, but instead varies with the applied reversebias voltage across the tuning diode over the temperature. The generalapproach of nullifying the temperature dependency of the tuning diode'sbuilt in contact potential φ by adding a forward bias diode ortransistor-emitter-follower in series with the tuning voltage of thetuning diode network comes at the cost of higher phase noise andnon-uniform thermal drift over the temperature range.

In accordance with an aspect of the present invention, the thermal driftis compensated for by introducing a common coupling-capacitor between aresonator, an active device and a tuning diode network of an oscillator.The coupling capacitor may comprise a slot-cut-microstripline or anyother variable capacitive storage element. The slot-cut-microstriplinecontrols the profile of the thermal drift and also acts as anevanescent-mode-buffer between the resonator and the active device, sothat the time average dynamic loaded Q of the resonator is enhanced andprovides low noise performance over the operating frequency band of theoscillator.

In particular, FIG. 2 shows an oscillator 200 in accordance with anaspect of the present invention. The oscillator includes athree-terminal device 210 having a first terminal 214, a second terminal216 and a third terminal 218. The three-terminal device may comprise anythree-terminal device that can provide a 180° phase shift between anytwo terminals and preferably includes a bipolar or field effecttransistor. A feedback-bias network 224 is connected between the firstand second terminals, 214, 216, respectively. Aslot-cut-printed-board-coupling network 230 is coupled to the secondterminal and to a tuning diode network 234. Theslot-cut-printed-board-coupling network 230 is also coupled to aresonator 240. In addition, the oscillator 200 includes a feedbackcapacitor 244 between the second and third terminals, 216, 218,respectively, and a pair of filters, 250, 252 coupled in series to thethird terminal 218. An output signal is taken between first filter 250and second filter 252.

In accordance with this aspect of the present invention, theslot-cut-printed-board-coupling network 230 compensates for capacitancechanges in the tuning-diode network 234 due to changes in operatingtemperature of the environment or the oscillator 200. In addition, andas discussed in further detail below, theslot-cut-printed-board-coupling network 230 may be implemented so as todefine the thermal drift profile of the oscillator, i.e., the change inoutput frequency due to change in operating temperature. The physicaldimensions of the slot-cut-printed-board may be chosen to define aparticular thermal profile, e.g., see FIGS. 5-8. Theslot-cut-printed-board-coupling network 230 also acts as an evanescentmode buffer between the resonator 240 and the three terminal device 210by storing additional energy that may develop in the oscillator as thetemperature changes. The additional energy is then typically releasedwithout increasing the phase noise of the output signal. In particular,the network 230 provides a storage element, e.g., a capacitor, thatgenerally operates to store excess energy that may develop in thecircuit due to temperature changes and releasing such energy so thatphase noise performance of the oscillator is controlled during thetemperature changes. For example, if the bias voltage increases due to achange in temperature, the capacitor assists in lowering the biasvoltage to or near the optimal operating point.

Turning now to FIG. 3, there is illustrated a oscillator 300 inaccordance with an aspect of the present invention. The oscillatorincludes an active device 310 having three terminals, 313, 315, 317. Theactive device 310 may comprise a bipolar transistor or field effecttransistor wherein the first, second and third terminals 313, 315, 317comprise, respectively, the collector, base and emitter nodes of thetransistor. In general, the active device 310 may comprise any devicethat can provide a 180° phase shift between the first terminal 313 andsecond terminal 315.

The first terminal 313 is connected to a feedback-bias network 323. Thenetwork 323 includes a voltage source Vcc coupled to the first terminal313 that is used for biasing the active device 310 by providing apredetermined voltage at the first terminal 313. The network 323 alsoincludes a pair of transistors Q2, Q3 (which are illustrated as bipolartransistors, but may also be field effect transistors) and associatedcircuit elements such as capacitors, resistors and inductors that couplea selected amount of the signal from the first terminal 313 to thesecond terminal 315.

The second terminal 315 is also capacitively coupled to tuning network329, slot-cut-printed-board-coupling capacitor 332 and a resonator 338.As shown, the tuning network 329, slot-cut-printed-board-couplingcapacitor 332 and resonator 338 are coupled in parallel. In addition,the tuning network 329 is capacitively coupled via coupling capacitor340. The slot-cut-printed-board-coupling capacitor 332 compensates forchanges in the capacitance, which are in turn caused by the changes inthe junction contact potential, e.g., dφ/dT, of the tuning network 329as a result of changes in the operating temperature of oscillator 300 orthe environment.

The oscillator 300 further includes a feedback capacitor 342 that iscoupled to the third terminal 317 through a resistor 344 and to groundthrough capacitor 348. Capacitor 342, resistor 344 and capacitor 348together form a network that feeds back a select portion of the signalfrom the third terminal 317 to the second terminal 315. The oscillator300 also includes a pair of filters 356, 358 coupled to the thirdterminal 317 that provide two-stage regenerative filtering. An outputsignal is capacitively coupled to output port 360 between the filters356, 358. As shown, filter 356 preferably comprises an LC filter andfilter 358 preferably comprises an RC filter. The time constants ofthese filters are preferably adjusted to the fundamental frequency ofoperation.

Turning now to FIG. 4, there is shown an oscillator 400 in accordancewith another aspect of the present invention. The oscillator 400includes a three-terminal device 410 that is inductively coupled to abias voltage source V_(cc) via first terminal 413. The second terminal415 of the device 410 is inductively coupled to a second voltage sourceV_(bb). A feedback capacitor C₁ is coupled to third terminal 417 througha resistor R. The third terminal 417 is also coupled to first and secondfilters, 422, 424, to provide regenerative filtering. In addition, theoscillator includes a slot-cut-microstrip-line-printed board 440 that iscoupled to a tuning diode network 442, a resonator 448 and the secondterminal 415 of the three terminal device 410. The tuning network 442includes circuit elements that are similarly arranged as discussed abovein relation to tuning network 329.

The resonator 448 is preferably a ceramic resonator and is capacitivelycoupled to terminal 452 of the slot-cut-microstrip-line-printed board440. The tuning network 442 and second terminal 415 are similarlycoupled to terminals 454 and 456 of the slot-cut-microstrip-line-printedboard 440. As shown, the slot-cut-microstrip-line-printed board 440includes a width, w, a height, h, and length dimensions, l₁ and l₂. Theboard 440 also includes a slot d that divides the base of the board 440into two regions defined by length dimensions, l₁ and l₂. Thesedimensions define the size of the board 440 and can be selected todefine the thermal profile of the oscillator. In accordance with thisaspect of the present invention, the structure is designed to increasethe loaded time average quality factor over the temperature range byselecting an optimum length-width ratio (L/W-ratio) of the each side ofthe slot-cut-microstrip-line coupling-capacitor. In general, the printedboard 440 preferably comprises a variable capacitor or storage elementthat operates as an evanescent mode buffer and allows a user to define athermal profile.

In particular, the L/W ratio and d may be selected so to provide athermal profile as shown in FIGS. 5-8. For example, as shown in FIG. 5the thermal profile 500 may designed to take the shape of a parabolaover the operating temperature range of −40° C. to 85° C. The dimensionsof the board 440 for providing a parabolic thermal profile as shown inFIG. 5 are as follows: l₁/w₁=1, l₂/w₂=0.5, d=0.01 inch, h=11 mils. Inaddition, l₁=0.06 inches, w₁=0.06 inches, l₂=0.03 inches, w₂=0.06 inchesand e_(r)=10. FIGS. 6-8 may be achieved by adjusting the ratios of l/w.Furthermore, by changing the dimensions of the board, different userdefinable profiles may be achieved. As shown, in FIG. 6 the thermalprofile 600 may take the shape of an inverted parabola. FIGS. 7 and 8illustrate linear thermal profiles 700, 800. In addition, as shown inFIGS. 7 and 8 the thermal drift is less than 100 kHz.

Turning now to FIG. 9, there is shown an oscillator 900 in accordancewith an aspect of the present invention. The oscillator 900 includessimilar circuitry to FIG. 5 except that the resonator 910 includes apair of ceramic resonators coupled in parallel with each other.

FIG. 10 shows a phase noise plot 1000 of an oscillator operating at 1200MHz in accordance with an aspect of the present invention. As FIG. 10shows, the phase noise is approximately −110 dBc/Hz at 1 kHz.

A voltage-controlled oscillator implemented in accordance with thepresent invention may be employed in any number of devices that are usedto communicate on data, telephone, cellular or, in general,communications network. Such devices may include but are not limited to,for example, cellular phones, personal digital assistants, modem cards,lap tops, satellite telephones. As a general matter, the oscillatorcircuitry shown in the various drawings and described above may beemployed in a PLL to either generate a clock signal that may be used totransmit or recover information transmitted or received over a network.In addition to wireless networks, the circuitry of the present inventionmay be employed in wired networks, satellite networks, etc.

In addition, and in accordance with additional aspects of the presentinvention, the slot-cut-microstrip-line board or coupling capacitor asdescribed above maybe further integrated with coupled resonatoroscillators disclosed in commonly assigned U.S. patent application Ser.Nos. 10/912,209 and 10/937,525, the disclosures of which areincorporated by reference herein.

Although the invention herein has been described with reference toparticular embodiments, it is to be understood that these embodimentsare merely illustrative of the principles and applications of thepresent invention. It is therefore to be understood that numerousmodifications may be made to the illustrative embodiments and that otherarrangements may be devised without departing from the spirit and scopeof the present invention as defined by the appended claims.

1. An oscillator, comprising: an active device having first, second andthird terminals; circuitry coupled between the first and secondterminals of said active device and operative to provide a bias voltageto said active device and feedback a select amount of phase noise intosaid active device; and a tuning diode coupled to the second terminal ofsaid active device through a slot-cut-printed-board coupling network. 2.The oscillator of claim 1, wherein the slot-cut-printed-board couplingnetwork acts as an evanescent mode buffer between a resonator coupledthereto and said active device.
 3. The oscillator of claim 1, whereinthe slot-cut-printed-board coupling network operates to control aprofile of the thermal drift of the oscillator.
 4. The oscillator ofclaim 1, further comprising a feedback capacitor coupled between thesecond and third terminals.
 5. The oscillator of claim 1, furthercomprising a first filter and a second filter coupled to the thirdterminal of said active device so as to provide two-stage regenerativefiltering.
 6. The oscillator of claim 5, further comprising meanscoupled between said first and second filters for providing an outputsignal.
 7. The oscillator of claim 1, wherein said active devicecomprises a field effect transistor and the first, second and thirdterminals respectively comprise the collector, base and emitter nodes ofthe transistor.
 8. The oscillator of claim 1, wherein said active devicecomprises a bipolar transistor and the first, second and third terminalsrespectively comprise the collector, base and emitter nodes of thetransistor.
 9. A voltage controlled oscillator, comprising: an activedevice; and circuitry coupled between a resonator, a tuning diodenetwork and the active device and operable to act as a common-couplingcapacitor between the resonator, the tuning diode network and saidactive device.
 10. The voltage controlled oscillator of claim 9, whereinsaid circuitry controls a thermal drift profile of the oscillator overan operating temperature range.
 11. The voltage controlled oscillator ofclaim 9, wherein said circuitry acts as an evanescent-mode-bufferbetween the resonator and said active device.
 12. The voltage controlledoscillator of claim 9, wherein the tuning diode network is capacitivelycoupled to said circuitry.
 13. The voltage controlled oscillator ofclaim 9, further comprising a network coupled to said active device forbiasing said three terminal device.
 14. The voltage controlledoscillator of claim 9, wherein the resonator comprises a ceramicresonator.
 15. The voltage-controlled oscillator of claim 91 wherein theresonator comprises a pair of ceramic resonators coupled in parallel.16. The voltage-controlled oscillator of claim 9, further comprising afirst filter and a second filter coupled to the active device so as toprovide two-stage regenerative filter at an output port of theoscillator.
 17. The voltage controlled oscillator of claim 9, whereinsaid active device comprises a field effect transistor.
 18. The voltagecontrolled oscillator of claim 9, wherein said active device comprises abipolar transistor.
 19. An apparatus, comprising: a phase lock loop forgenerating a clock signal used to transmit or recover informationcommunicated from or to the apparatus, wherein the phase lock loopincludes a voltage-controlled oscillator for generating the clocksignal, the voltage controlled oscillator comprising, an active device;and a slot-cut-microstrip-line coupled between a resonator, a tuningdiode network and the active device and operable to act as acommon-coupling capacitor between the resonator, the tuning diodenetwork and said active device.
 20. The apparatus of claim 19, whereinthe apparatus comprises a wireless device.
 21. The apparatus of claim19, wherein the wireless device is a cellular telephone.
 22. Theapparatus of claim 19, wherein the apparatus comprises a personaldigital assistant.
 23. A telephone, comprising: a phase lock loop forgenerating a clock signal used to transmit or recover informationcommunicated from or to the telephone, wherein the phase lock loopincludes a voltage-controlled oscillator for generating the clocksignal, the voltage controlled oscillator comprising, an active device;and circuitry coupled between a resonator, a tuning diode network andthe active device and operable to act as a common-coupling capacitorbetween the resonator, the tuning diode network and said active device.24. The telephone of claim 23, wherein the information is communicatedover a wireless network.
 25. The telephone of claim 23, wherein theinformation is communicated over a wired network.
 26. A method ofdefining a thermal profile of an oscillator, comprising coupling acapacitor between a resonator, a tuning diode network and an activedevice; and operating the capacitor as an evanescent mode buffer betweenthe resonator and the active device to compensate for drifts in anoutput frequency of the oscillator due to temperature changes.
 27. Themethod of claim 26, further comprising biasing the active device atpredetermined voltage such that the capacitor maintains thepredetermined voltage level to compensate for drifts in an outputfrequency of the oscillator due to temperature changes.
 28. The methodof claim 26, wherein the evanescent mode buffer compensates for driftsin the output frequency of the oscillator due to temperature changes bystoring additional energy generated by the oscillator due to temperaturechanges.